Transmitarray antenna

A transmitarray antenna (or just transmitarray or called as layered lens antenna ) is a phase-shifting surface (PSS), a structure capable of focusing electromagnetic radiation from a source antenna to produce a high-gain beam. Transmitarrays consist of an array of unit cells placed above a source (feeding) antenna. Phase shifts are applied to the unit cells, between elements on the receive and transmit surfaces, to focus the incident wavefronts from the feeding antenna. These thin surfaces can be used instead of a dielectric lens. Unlike phased arrays, transmitarrays do not require a feed network, so losses can be greatly reduced. Similarly, they have an advantage over reflectarrays in that feed blockage is avoided. It is worth clarifying that transmitarrays can be used in both transmit and receive modes: the waves are transmitted through the structure in either direction. An important parameter in transmitarray design is the F/D ratio, which determines the aperture efficiency. F is the focal length and D is the diameter of the transmitarray. The projected area of the feeding antenna determines the illumination efficiency of a transmitarray panel. Provided that the insertion loss of each unit cell is minimised, an aperture area appropriate to the feed radiation pattern can efficiently focus the wavefronts from the feed.

A transmitarray antenna (or just transmitarray or called as layered lens antenna ) is a phase-shifting surface (PSS), a structure capable of focusing electromagnetic radiation from a source antenna to produce a high-gain beam. Transmitarrays consist of an array of unit cells placed above a source (feeding) antenna. Phase shifts are applied to the unit cells, between elements on the receive and transmit surfaces, to focus the incident wavefronts from the feeding antenna. These thin surfaces can be used instead of a dielectric lens. Unlike phased arrays, transmitarrays do not require a feed network, so losses can be greatly reduced. Similarly, they have an advantage over reflectarrays in that feed blockage is avoided. It is worth clarifying that transmitarrays can be used in both transmit and receive modes: the waves are transmitted through the structure in either direction. An important parameter in transmitarray design is the F/D ratio, which determines the aperture efficiency. F is the focal length and D is the diameter of the transmitarray. The projected area of the feeding antenna determines the illumination efficiency of a transmitarray panel. Provided that the insertion loss of each unit cell is minimised, an aperture area appropriate to the feed radiation pattern can efficiently focus the wavefronts from the feed. First, consider fixed transmitarrays. At each location on the surface of the structure, the unit cells are physically scaled or rotated in order to obtain the required amplitude and phase distribution. Thus, only one focusing direction is available. The aim is to approximate the ideal phase distribution, such as \Delta \phi(x,y) = \frac{-2\pi}{\lambda_{0}}\sqrt{x^2 + y^2 + F^2} for a feed located at (0,0,-F) , which can be achieved by discretising the surface of the transmitarray into several Fresnel zones. High aperture efficiency (55%) can be achieved at oblique angles of incidence using precision-machined double split ring slot unit cells. A switched-beam transmitarray covering the 57 – 66 GHz band has been reported. Three different types of unit cells were used, based on patches and coupling slots. Similarly, a 60 GHz design used unit cells with a 2-bit phase resolution and selected an optimal F/D ratio to widen the bandwidth. When F/D = 0.5, a scan loss of 2.2 dB was achieved at a 30° steering angle. Different types of unit cells have been used within the same transmitarray. In, slot elements were placed near the centre of the transmitarray, as their polarisation performance is better at normal incidence, whereas double square ring slot elements were used at the edges, as they perform better at oblique incidence angles. This enabled the subtended (flare) angle of the feed horn to be increased, and hence the length of the horn, and the overall antenna size, to be reduced. Unit cells were not required at the centre of the transmitarray, where the phase shift was 0°. This reduced the insertion loss to around 1 dB at 105 GHz, as the majority of the beam amplitude was in the central region. In a different design, substrate integrated waveguide (SIW) aperture coupling was employed to reduce insertion losses and widen the bandwidth of a transmitarray operating at 140 GHz. Due to the large number of vias required, this performance improvement was at the expense of a more complex and costly fabrication. It has been shown that transmitarray implementation can be divided into two approaches: layered-scatterer and guided-wave. The first approach uses multiple coupled layers to achieve a phase shift, but has poor sidelobe level (SLL) performance when steering due to higher-order Floquet modes. The second approach enables wider steering, at the expense of increased hardware cost and complexity. \phi_{m}(x_{m}, y_{m}, z_{m}) = -k_{0}(\sin{\theta_{0}}\cos{\phi_{0}}x_{m} + \sin{\theta_{0}}\sin{\phi_{0}}y_{m} + \cos{\theta_{0}}z_{m}) where ( \theta_{0} , \phi_{0} ) are the elevation and azimuth steering directions, and (x_{m}, y_{m}, z_{m}) are the coordinates of unit cell m . Note that x_{m} = \left(m + \frac{M-1}{2}\right)d , y_{n} = \left(n + \frac{N-1}{2}\right)d , and z_{m} = 0 . M and N are the total numbers of unit cells in the x - and y -directions respectively. When steering in azimuth only, this simplifies to: \phi_{m}(x_{m}, y_{m}) = k_{0}(d_{m} - \sin{\theta_{0}}(x_{m}\cos{\phi_{0}} + y_{m}\sin{\phi_{0}})) where d_{m} = \sqrt{(x_{m} - x_{f})^2 + (y_{m} - y_{f})^2 + z_{f})^2} and ( x_{f} , y_{f} , z_{f} ) are the coordinates of the feed, in this case (0,0,- F ). The overall radiation pattern can be calculated, using. Here, terms are combined to express the formula in full: E(\theta, \phi) = \sum_{m=1}^{M} \sum_{n=1}^{N} \cos^{q_{e}}\left({\theta - \theta_{0}}\right) \frac{\cos^{q_{f}}\left({\theta_{f mn}}\right)}{\sqrt{(md)^2 + (nd)^2 + F^2}}|T_{mn}| e^{j \Psi_{mn}}\times e^{-jk\left(\sqrt{(md)^2 + (nd)^2 + F^2} - d(m\sin{\theta}\cos{\phi} + n\sin{\theta}\sin{\phi})\right)} where the radiation pattern of the steered array source is modelled as \cos^{q_{e}}\left({\theta - \theta_{0}}\right) . The term e^{j \Psi_{mn}} corresponds to the phases applied to the transmitarray unit cells, to undo the phase variation due to the geometry of the cells from the feed, i.e. e^{j \Psi_{mn}}e^{-jk \angle G_{mn}} = 1 . G_{\textrm{edge}} = 10\log_{10}\left(\cos^{n}{\theta_{sub}}\right) An edge taper of around -10 dB is desired, so that the illumination efficiency is maximised. For a planar (conventional) transmitarray, fed by an antenna with radiation pattern G_{f}(\theta,\phi) = \cos^{n}{\theta} , and subtended angle \theta_{sub} = \tan^{-1}\left({\frac{D}{2F}}\right) , the taper efficiency is calculated by: \eta_{s} = 1 - \cos^{n+1}{\theta_{sub}} \eta_{t} = \frac{2n}{\tan^2{\theta_{sub}}} \frac{(1 - \cos^{(n/2) - 1}{\theta_{sub}})^2}{(\frac{n}{2} - 1)^2 (1 - \cos^{n}{\theta_{sub}})} \theta_{sub} is a function of F/D . Note that \cos(\tan^{-1}{x}) = \frac{1}{\sqrt{1+x^2}} , so using x = \frac{D}{2F} , this formula can be expressed in terms of F/D , rather than the subtended angle. The illumination efficiency is the product of these: \eta_{i} = \eta_{s}\eta_{t} . The overall aperture efficiency \eta_{ap} is obtained by multiplying by material losses and any directivity reduction terms. Consider the structure of the proposed 1-bit unit cell, which operates at 28 GHz. It is based on the design presented in. It consists of two metal layers, printed on a Rogers RT5880 substrate material having a thickness of 0.254 mm, a dielectric constant of 2.2, and a loss tangent of 0.0009. Each metal layer consists of a pair of crossed slots, and the incident fields are vertically polarised ( E_{y} ). By selecting a symmetrical unit cell shape, they can be adapted for dual linear or circular polarisation. The two metal layers are separated by a 3 mm thick layer of ePTFE material (of dielectric constant \epsilon_{r} = 1.4), which creates a 100° phase shift between these layers. The unit cell has reduced thickness and insertion loss compared with multilayer designs. The unit cell can be reconfigured between two phase states, OFF (0°) and ON (180°). For the OFF state, it has a Jerusalem cross slot structure. In the ON state, the slots are not loaded with Jerusalem cross (JC) shaped caps, producing a large phase change. Due to the use of single-pole resonators (a two-layer structure), the transmission performance was challenging to achieve, requiring fine-tuning of the unit cell physical dimensions. Both unit cell states were simulated in CST Microwave Studio using Floquet ports and the frequency domain solver. This included the magnitude and phase of the E_{y} transmission coefficient through the unit cell in ON and OFF states. A phase change of 189° was observed, which is close to 180°, and the transmission magnitude is at least -1.76 dB at 28 GHz for both states. For the JC cells, the surface currents are in opposite directions (anti-phase) on each conductor layers, whereas for the CS cells, the surface currents are in the same direction (in-phase). The phase difference between states is given by: \Delta\phi = \angle{S_{21}}_{\textrm{OFF}} -\angle{S_{21}}_{\textrm{ON}} . The MACOM MA4GP907 diode has an ON resistance R_{\textrm{ON}} = 4.2 \Omega , an OFF resistance R_{\textrm{OFF}} = 300 k \Omega , and small parasitic inductance and capacitance values ( L_{\textrm{ON}} = 0.05 nH, C_{\textrm{OFF}} = 42 fF in the 28 GHz band). Given that it has a high OFF resistance value, and that the switching time is very fast (2 ns), this component is suitable for the design. The position and orientation of the bias lines must be chosen to minimise their effect on the transmission of the incident waves through the structure. If the lines are sufficiently narrow (width up to 0.1 mm), they will present a high impedance, so will have less effect on the wavefronts. As they act as a polarising grid, the bias lines should be perpendicular to the incident field direction. This design has no ground plane, so each group of active unit cells must have both a V_{\textrm{bias}} and a ground connection. As groups of cells share the same bias voltages, these lines can be routed between adjacent cells. The required number of external control lines is equal to the number of beam directions supported, so is inversely proportional to the steering resolution. The bias lines could be implemented as large blocks of copper around the unit cells, separated by thin gaps (through which the RF wave propagation is heavily attenuated). The gaps may need to be meandered to form DC block capacitors. Radial stubs or high-impedance lines of length \frac{\lambda_{g}}{4} (a quarter of a guided wavelength) could be used as chokes (inductors) on the external control lines, to prevent the RF signal from affecting the DC control circuitry.

Text derived from https://en.wikipedia.org/w/index.php?oldid=1355442511 under CC BY-SA 4.0; modifications have been made.